This application claims the priority of German Application 19961798.8, filed Dec. 22, 1999, the disclosure of which is expressly incorporated by reference herein.
The invention relates to a method and an arrangement for regulating the phase current in a switched reluctance machine, whose stator windings in each phase are each connected to a DC chopper controller which is connected to a regulator which processes the control error between the required current value and the measured actual current value and applies pulse-width-modulated electrical pulses to the DC chopper controller.
An arrangement of the type described above is known (U.S. Pat. No. 5,754,024). The DC chopper controller in each phase of the known arrangement comprises a first series circuit of a switching transistor with a freewheeling diode, and a second series circuit of a freewheeling diode with a switching transistor. The switching transistor in the first series circuit is connected to the positive pole of a DC voltage source, and the switching transistor in the second series circuit is connected to the negative pole of the DC voltage source. The freewheeling diodes are reverse-biassed with respect to the polarity of the DC voltage source. The control electrodes of the switching transistors, which are IGBTs, are connected to a pulse-width modulator which has a first input connected to a clock generator, a second input connected to a comparator, and a third input to which an on/off signal is applied. The phase winding is arranged in series with a current sensor between the points where the switching transistors are connected to the freewheeling diodes. A first input of the comparator has a required current value applied to it, and a second input has the actual current value from the current sensor applied to it. The required current value and the on signal together with the off signal for the pulse-width modulator are determined as a function of the rotor position, measured by a sensor. The pulse-width modulator starts when it is intended to apply current to the respective winding, and stops when it is intended to stop current flowing in the winding once again.
German Patent DE 43 10 772 C2 discloses a method for regulating the phase current in a switched reluctance machine, whose stator windings in each phase are each connected to a DC chopper controller, which is connected to a regulator which processes the control error between the required current value and the measured actual current value and applies pulse-width-modulated electrical pulses to the DC chopper controller. In the case of the control circuit disclosed there, the control error between the required current value and the actual current value is supplied to a PI regulator.
European Patent EP 0 684 693 A2 discloses an arrangement for regulating the phase current of brushless DC machines and switched reluctance machines, in which the control error is determined from the required values and from actual current values obtained by sampling and equidistant intervals.
A three-point regulator with hysteresis is suitable for regulating the phase current in the reluctance machine. The output of the three-point regulator can assume three states, each of which can be associated with a switching state of a converter or DC chopper controller. The association with the xe2x80x9con, short-circuitxe2x80x9d and xe2x80x9coffxe2x80x9d switching states of the current regulator allows the phase current to be regulated not only in motor operation but also in generator operation down to zero speed, without the three-point regulator needing to be switched. If the three-point regulator has identical switching thresholds when the reluctance machine is being operated as a motor and as a generator, this, in fact, results in a higher mean current value in generator operation than in motor operation. This effect can be minimized by hysteresis loops which are shifted one above the other. One advantage of a three-point regulator with hysteresis is its simple structure.
A disadvantage of the three-point regulator is that the converter switching frequency caused by the three-point regulator depends not only on the switching thresholds but also on the rate of current change in the machine winding, which in turn depends on the phase voltage, the winding resistance, the present current value, the phase inductance (which is dependent on the rotor position) and the rotation speed. Taking account of these influencing variables, the switching thresholds of the three-point regulator must be selected such that the maximum switching frequency of the power semiconductors in the converter is not exceeded. During operation of the reluctance machine, this results in switching frequencies which are well below the maximum switching frequency and are in the audible range. As a result the reluctance machine produces irritating noises.
The invention is based on the problem of specifying a method which can be matched flexibly to different situations that occur with reluctance machines, and an arrangement for regulating the current in phase windings of a switched reluctance machine, in which irritating noise from the reluctance machine, caused by the switching frequencies of the converter active devices is largely avoided and in which the phase currents can be set dynamically and quickly to the predetermined required values.
According to the invention, with regard to a method of the type described initially, the problem is solved by determining the control error from the required values and from actual current values obtained by sampling at equidistant intervals. Also a first manipulated variable is formed from the control error digitally using a proportional-integral characteristic, by linear superimposition of an integral element and a proportional element which is multiplied by the respective electrical angular position of the reluctance machine. Furthermore the first manipulated variable has a second manipulated variable superimposed on it linearly, which is formed as a pilot control value of a characteristic value by multiplication by the rotation speed, which characteristic value is read, as a function of the phase current and as a function of the electrical angular position of the rotor, from a characteristic map, which includes the derivative of the magnetic flux of the reluctance machine with regard to the electrical angular position, as a function of the electrical angular position of the rotor of the reluctance machine and as a function of the phase current. The method according to the invention allows the phase currents to be well regulated even at high rotation speeds and at high pulse-width-modulation frequencies, as well as allows for rapid changes in the induced phase voltage.
One preferred embodiment provides that characteristic values are stored in a table as a function of the electrical rotor angle positions. The characteristic values are determined from a data set with the magnetic flux values of the reluctance machine as a function of the electrical rotor angular position and of the phase currents by deriving the flux values with respect to the rotor angle, by division by a saturation current which is typical for the transition to the saturated magnetic state, and by forming the mean values of the respective rotor position. The pilot value is formed by multiplication of the characteristic value, which is read as a function of the measured electrical rotor angular position, by the rotation speed and the phase current. In this embodiment, relatively little memory capacity is required for storing the characteristic values. The approximate determination of the rotational voltage value for the pilot control is not a disadvantage, because the regulator can quickly compensate for a relatively small error between the required value and the actual value.
In one expedient embodiment, the control error at the time tK=k*TA is calculated using the following equation e (k)=w (k)xe2x88x92x (k) where e is the control error, W is the required current value, x is the actual current value, tK is the time, k is the number of sampling intervals and TA is the sampling time, and in that the manipulated variable is calculated using the following equation:
y(k)=Kp*e(k)+YI(kxe2x88x921)+KI*e(kxe2x88x921), 
where y (k) is the manipulated variable, Kp is the proportional gain, YI is the integral element of the manipulated variable, KI is the product of the proportional gain and the quotient of the sampling time and the readjustment time of the regulation, and e (k) is the control error. The method described above allows the manipulated variable to be determined in a relatively short time from the control error. The regulator computation time is thus very short. Computation time in this case refers to the time which passes from reading the actual value via an A/D converter to the time at which the control signal is applied to the converter.
In particular, the readjustment time of the PI regulation is on the one hand set to the time constant of the phase winding of the reluctance machine, and the factor of the integral element is on the other hand set using the following relationship:       K    1    =            T      A              2      ⁢              K        S            *              T        I            
where KI is the factor for the integral element, TA is the sampling interval, Ks is the path time constant of the controlled system, and Tt is the dead time of the regulation, while the gain factor is readjusted as a function of the rotor position using the following relationship:             K      p        ⁡          (      γ      )        =                    T        1            ⁡              (        γ        )                    2      ⁢              K        S            *              T        t            
where Kp (xcex3) is the gain factor, T1 (xcex3) is the current-dependent and rotor-position-dependent time constant of the phase winding, Ks is the path time constant and Tt is the dead time of the regulation. Such a setting process results in the control loop having a good time response.
Switched reluctance machines have a time constant which is dependent on the current and the rotor position and for which:             T      1        ⁡          (              γ        ,        i            )        =            Ψ      ⁡              (                  γ          ,          i                )                    i      ,              *        R            
where T1 is the time constant, xcexa8 is the magnetic flux, xcex3 is the rotor position, i the phase current and R the pure resistance of the phase winding.
It is particularly advantageous if the time constant T1 (xcex3) of the phase winding for the q-position and for the d-position of the rotor is determined using the following relationships:                               T                      1            ⁢            q                          =                              Ψ            ⁡                          (                                                γ                  q                                ,                                  i                  max                                            )                                                          i              max                        *            R                                                            T                      1            ⁢            d                          =                              Ψ            ⁡                          (                                                γ                  d                                ,                                  i                  max                                            )                                                          i              max                        *            R                              
and, for the intermediate positions of the rotor between the q-position and the d-position, are multiplied by the product of the electrical angular position of the rotor and the ratio T1d/T1q, where T1q is the time constant of the phase winding in the q-position of the rotor, T1d is the time constant of the phase winding in the d-position of the rotor, xcexa8 (xcex3q, imax) is the magnetic flux of the reluctance machine when the rotor is in the q-position and the current is a maximum during operation of the reluctance machine, R is the pure resistance of the phase winding and xcexa8 (xcex3d, imax) is the magnetic flux of the reluctance machine when the rotor is in the d-position and the current is at the maximum value at which the reluctance machine is intended to operate. This method allows the time constant of the phase winding to be determined with sufficient accuracy with a short computation time.
In a further preferred embodiment, whenever a switch-on angle for the phase current is reached, a start pulse is produced by the converter using the following relationship:       PWM    start    =            (                        n                      n            max                          +                              i            w                                I            max                              )        *          PWM              100        ⁢        %            
where PWMstart is the start signal, n is the measured rotation speed, nmax is the maximum rotation speed, Imax is the maximum current of the drive, iw is the required current setting and PWM100% is the pulse-width pulse for full control. Using this start value, the sampling clock rate of the regulation and the process of switching the phases on and off are coordinated such that no angular errors result from the asynchronous relationship between the sampling clock rate and the switching of the winding phases, which is dependent on the rotation speed. Furthermore, the reaction time of the regulation is minimized. In addition, this avoids any discontinuities in the transition from pulsed operation of the reluctance machine to block operation.
In an arrangement for regulating the phase current in a switched reluctance machine, whose stator windings in each phase are each connected to a DC chopper controller, which is connected to a regulator which processes the control error between the required current value and the measured actual current value and applies pulse-width-modulated electrical pulses to the DC chopper controller, the problem is solved, according to the invention, in that the regulator has a microcontroller to whose input side required current values and actual current values can be supplied via an A/D converter, and to which rotation position signals can be supplied which are produced by a rotation position sensor in the reluctance machine. A program calculator the control error and the manipulated variable, using a PI characteristic, is stored in the regulator. The program has a part for separately calculating the proportional and I-elements in accordance with the PI characteristic. The proportional and I elements are added. A constant is stored for calculating the I-element as the quotient of a constant sampling interval and the product of twice the path gain of the controlled system and the dead time of the regulator. In order to determine the gain factor, values of the time constant of the phase winding are stored as a function of the rotor position in a memory. Furthermore, in order to determine pilot values (which can be formed by multiplication of the rotor rotation speed by characteristic values and are superimposed on the manipulated variable of the output of the regulator), a characteristic map, which contains the derivative of the magnetic flux of the reluctance machine with respect to the electrical angular position as a function of the electrical rotor angle position and the phase current, is stored as a function of the phase current and of the rotor angular position.
A considerable saving in memory space is achieved if the characteristic map includes a series of characteristic values which have been determined in the following way: differentiation of the magnetic flux values of the reluctance machine as a function of the electrical rotor angular position and of the phase currents with respect to the rotor angular position; division of the differentiated values by a saturation current which is typical for the transition to the saturated magnetic state; and formation of the mean values for the respective rotor position.
The invention will be described in more detail in the following text with reference to an exemplary embodiment which is illustrated in the drawings and from which further details, features and advantages are evident.
Other objects, advantages and novel features of the present invention will become apparent from the following detailed description of the invention when considered in conjunction with the accompanying drawings.